Voltage compensating CMOS input buffer

Casper, Stephen L.;

The CMOS voltage compensating input buffer circuit of the present invention provides a means to stabilize input level trip points and is comprised of a voltage compensating circuit having an input node and an output drive node coupled to an input buffer. The voltage compensating circuit receives its input from a voltage adjusting circuit that follows changes in V.sub.CC while its output drive node is coupled to the series connected CMOS input buffer circuit having an input node and an output node. The buffer's input node receives a signal that VIH/VIL trip points will determine if the output is to be a high or a low and the buffer's output node then couples the resultant level to an output buffer circuit comprised of a CMOS inverter which provides the final output drive. The present invention provides trip point levels corresponding to industry standard VIH/VIL levels to accurately determine the corresponding output with operating voltage supplies (regulated or unregulated) operating between 2 V to 7.5 V.

FIELD OF THE INVENTION

This invention relates to a CMOS voltage compensating input buffer circuit formed in a semiconductor integrated circuit (IC) and in particular is a circuit that may be used in various memory devices, such as dynamic random access memories (DRAMs) or erasable programmable read only memory families (EPROMs, EEPROMs, etc.).

BACKGROUND OF THE INVENTION

Standard IC operating input levels (VIH and VIL) have been set by industry standards whether the given IC is operating with a regulated power supply with very tight tolerances or with an unregulated supply. Operating input levels for a typical DRAM for example, are set at 0.8 V+/-100 mv for VIL and 2.4 V+/-100 mv for VIH. Typically, DRAMs are operating from unregulated supplies that may vary from 4.5 V to 5.5 V and the input level standards must still be adhered to. There is also may instances wherein qualification of DRAMs to meet industry reliability standards requires operating the DRAM at much higher voltages than the 5.5 V specification (i.e. 7.5 V for burn-in testing of a part's mortality rate) or lower than 4.5 V (i.e. 3 V supplied ICs). In order to meet this criteria, the DRAM's input buffers must correctly accept the various signals that are used to operate a DRAM (such as the signals used to read and write data), must be able to differentiate precisely between an intended low input(determined by VIL) to that of an intended high signal (determined by VIH).

SUMMARY OF THE INVENTION

The CMOS voltage compensating input buffer circuit of the present invention provides a means to stabilize input level trip points, and is comprised of a voltage adjusting circuit providing a voltage adjusting output node coupled to a voltage compensating circuit.

The voltage compensating circuit's drive node is coupled to a series connected CMOS input buffer circuit having an input node and an output node. The input node receives a signal that VIH/VIL trip points which determine if the output is to be a high or a low. The output node then couples the resultant level to an output buffer circuit comprised of a CMOS inverter which provides the final output node drive.

In order to maintain device reliability for future IC generations with sub-micron devices, the present invention provides trip point levels corresponding to industry standard VIH/VIL levels to accurately determine the corresponding output with operating voltage supplies operating between 2 V to 7.5 V. In fact, the present invention will work with any operating voltage supplies that will be used to power future ICs. The present invention also allows the use of both regulated or unregulated power supplies as trip point levels for VIH/VIL are accurately maintained with fluctuating voltages.

BRIEF DESCRIPTION OF THE DRAWINGS

FIG. 1 is a schematic of a CMOS voltage input buffer, which may be viewed as related art;

FIG. 2 is a schematic of the preferred embodiment depicting a voltage compensating CMOS input buffer;

FIG. 3 is a computer simulation comparing the circuit response of FIG. 1 to the present invention (shown in FIG. 2) during circuit operation at VCC=6 V; and

FIG. 4 is a computer simulation comparing the circuit response of FIG. 1 to the present invention (shown in FIG. 2 during circuit operation at VCC=4 V .

DETAILED DESCRIPTION OF THE PREFERRED EMBODIMENT

FIG. 2 depicts a schematical representation of the preferred embodiment for the input buffer voltage compensating circuit with the present invention comprising, voltage adjusting circuit stage 21, voltage compensating stage 22, input stage 23 and output stage 24. The voltage adjusting circuit stage 21 and output stage 24 are stages used to interface between the present invention and outer lying circuitry. Though these circuits are used to discuss the present invention they can be any circuit of designer's choice that provides these functions. For example, output stage 24 need not be present as the output could simply be taken from the output of input stage 23 without departing from the scope of the present invention.

Voltage adjusting circuit 21 comprises PMOS transistors Q1 and Q2 connected in series fashion with the source of Q1 coupled to V.sub.CC and its gate and drain terminals coupled together as well as coupled to the source of Q2. Q2 also has its gate and drain terminals coupled together as well as coupled to the drain of NMOS transistor Q3 (referred to as node A). Q3 has its gate terminal coupled to V.sub.CC and its source terminal coupled to V.sub.SS (or ground). Voltage adjusting circuit 21 is the preferred voltage adjusting circuitry of the present invention, however any circuit that adjusts the voltage at node A can be used for stage 21 without departing form the operational scope of the present invention.

Voltage compensating circuit 22 comprises PMOS transistor Q4 with its source terminal coupled to V.sub.CC1 and its gate terminal coupled to node A of voltage adjusting stage 21. The significance of Q4, will be brought to light later in this document when comparisons of circuit operation simulations between a conventional input buffer depicted in FIG. 1 and the present invention depicted in FIG. 2 are made.

Input stage 23 is comprised of a CMOS inverter, formed from PMOS transistors Q5 and Q6 and NMOS transistors Q7 and Q8. The source of Q5 is couple to the drain of voltage compensating transistor Q4, while its drain terminal is coupled to the source terminal of Q6. The drain of Q6 is in turn coupled to the drain of NMOS transistor Q7, thereby providing an output node. The source of Q7 is coupled to the drain of Q8 while the source of Q8 is coupled to V.sub.SS. The gates of all input stage transistors Q5, Q6, Q7 and Q8 are coupled together to form the input node.

Output stage 24 is comprised of CMOS inverter U1 with the input terminal of U1 coupled to drain connections of input stage transistors Q6 and Q7, while its output terminal becomes the output node (Out2).

• Video bit rate control method Attenuated lentivirus vectors expressing interferon
• Electrical connector Crosslinkable acrylonitrile copolymers
• Post anchor Polynucleotide sequencing
• Outerwear and bag in one Cutting tool with clamping device
• Mirror with internal support plate Analog to digital converters
• IC card for image processor Nurse's mitt
• Surface mounted device parts counter Certain 1-alkyl-2-dichlorophenyliminopyrrolidines
• Reflection liquid crystal display element Double wall steel tank
• Esterification process Projection lens system and apparatus
• Bath cabinet and hinge therefor Multimedia search system
• Baseball bat Secure optical multiplex communication system
• Storage case Flexible two-sided multiple-sport goal
• 1-((A-substituted phenyl-.omega.-substituted piperazinyl)alkenyl) cyclohexanol Camera with motor drive
• Continuous tension exerciser High density information record lubricants
• Thermal printing system Optical projectors
• Shoe rack Moisture-curing polyurethane hot-melt adhesive
• Magnetic bubble device Process for producing polyurea resin
• Controllable conduction device Modified deoiling-dewaxing process
• Combination interpolator and distance divider Autofocusing system for a camera
• Slicing machine for cold cuts Fluid measuring probe
• Fiber optic connector Diaper-changing aid
• Salat time alarm electronic timepiece Combustion engine air supply system
• Implantable treatment material device Semiconductor device
• Cooling fan control Grain drying and storage apparatus
• Multi-display mode picture frame Alternatively spliced circulating tissue factor
• .beta.-amylase gene Hydrocarbon treating catalyst
• Magnetic-tape storage apparatus In-case bottle filling apparatus
• Speaker diaphragm Optical fiber grating packages
• 2-Amino-3-(3'-hydroxy-phenyl)-propanols and salts thereof Quasi-zoom lens optical focus sensor
• Process for processing granules Linear sorting machine and method
• Two-piece bottle carrier

For a general understanding of circuit operation assume for sake of illustration that the threshold voltage (vt.) for all n-channel devices is approximately equal to 0.8 V and -0.8 V for all p-channel devices (typical threshold voltages range from 0.6 V to 1.2 V). Further assume that series transistors in their respective stages are matched. It is also important to note that V.sub.CC1 and V.sub.SS need not be the IC's operating supplies, (i.e. normally referred to as power and ground), but instead they may represent voltage potentials in a circuit which provides a difference in potential (preferably a potential between 2 V to 7.5 V). For example, V.sub.CC1 may equal 15 V and V.sub.SS may equal 7.5 V or simply V.sub.CC1 may result by having an enabling transistor connected between the source of Q4 and V.sub.CC while V.sub.SS is at ground.

GENERAL OPERATION

Referring to back to FIG. 2, voltage adjusting circuit 21 may be any circuitry that will adjust node A according to the operating level of V.sub.CC (preferably V.sub.CC is between the range of 2 to 7.5 V, but due to the addition of voltage compensating transistor Q4, it is conceivable that the present invention would provide stable trip point levels for VIH at V.sub.CC levels above 7.5 V and maintain trip point levels for VIL at V.sub.CC below 2 V). As V.sub.CC rises, voltage adjusting circuit increases the voltage potential at node A thereby reducing the drive of Q.sub.4 (of circuit 22) which will reduce the voltage potential at node B. This lowered potential at node B allows the inverter configuration of input stage 23 to operate safely in the set VIH/ VIL levels of 2.4 and 0.8 V. As operating comparisons between FIGS. 1 and 2 will show, Q4 maintains VIH trip point levels as V.sub.CC increases, as VIH trip points tend to rise as the threshold voltage of operating transistors increase with V.sub.CC. In other words, worse case for VIH is at higher V.sub.CC operating levels. Q4 on the other hand, maintains VIH/VIL trip point levels for low V.sub.CC (i.e. 4 V and below). Also, depending on designer's choice, Q4 could be an NMOS transistor and instead connect between the source of Q8 and V.sub.SS which would in effect raise Q8's source above V.sub.SS, thereby maintaining VIH/VIL trip point levels for increases in V.sub.CC.

As V.sub.CC decreases, voltage adjusting circuit decreases the voltage potential at node A, thereby increasing the drive of Q.sub.4 (of circuit 22) which will increase the voltage potential at node B closer to V.sub.CC. This higher potential at node B allows the inverter configuration of input stage 23 to operate safely in the set VIH/VIL levels of 2.4 and 0.8 V.

The circuit operation of the present invention in light of changing V.sub.CC operating levels, will become evident in the representative cases of an input signal transitioning between VIH (2.4 V) and VIL (0.8 V) as compared hereinafter, between the operation of a conventional input buffer and the present invention.

VIH TO VIL TRANSITIONS (V.sub.CC =V.sub.CC1 =6 V)

In this discussion, refer to the circuit simulations, as graphed in FIG. 3, which compare the operations of the circuits of FIGS. 1 and 2. As FIG. 3 shows, both input buffers of FIGS. 1 and 2 are operating from a power supply potential of 6 V. Both output nodes, Out1 (of FIG. 1) and Out2 (of FIG. 2), are at an output potential level of 6 V with the inputs set at a beginning VIH potential level of 2.4 V at T=0 ns. Looking only at ideal transition curves, Out1 transitions from 6 V down to 0 V between 68 and 78 ns when the input level approaches approximately 2.0 V. Out2, on the other hand, transitions from 6 V down to 0 V between 85 and 95 ns when the input level approaches approximately 1.8 V. The present invention adds an additional VIH/VIL guardband of 0.2 V while guaranteeing a high level (6 V in this case) at Out2.

VIL TO VIH TRANSITIONS (V.sub.CC =V.sub.CC1 =6 V)

Continuing with the simulations of FIG. 3, the input level approaches a VIL level of 0.8 V and then transitions toward the VIH level of 2.4 V. Out2 (now at 0 V) transitions to 6 V between 415 and 425 ns once the input level approaches 1.8 V. Out1 (also at 0 V), on the other hand, does not transition to 6 V until between 430 and 440 ns when the input level approaches 2.0 V. As before, a 0.2 V guardband to VIH/VIL is demonstrated.

VIH TO VIL TRANSITIONS (V.sub.CC =V.sub.CC1 =4 V)

In this discussion, refer to the circuit simulations, as graphed in FIG. 4, which compare the operations of the circuits of FIGS. 1 and 2. As FIG. 4 shows, both input buffers of FIGS. 1 and 2 are operating from a power supply potential of 4 V. Both output nodes, Out1 (of FIG. 1) and Out2 (of FIG. 2), are at an output potential level of 4 V with the inputs set at a beginning VIH potential level of 2.4 V at T=0 ns. Looking only at ideal transition curves, Out1 transitions from 4 V down to 0 V between 158 and 163 ns when the input level approaches approximately 1.35 V. Out2, on the other hand, transitions from 4 V down to 0 V between 155 and 160 ns when the input level approaches approximately 1.4 V. This simulation shows that the present invention maintains VIH/VIL trip point levels thereby guaranteeing a low level (0 V in this case) at Out2.

VIL TO VIH TRANSITIONS (V.sub.CC =V.sub.CC1 =4 V)

Continuing with the simulations of FIG. 4, the input level approaches a VIL level of 0.8 V and then transitions toward the VIH level of 2.4 V. Out2 (now at 0 V) transitions to 4 V between 351 and 357 ns once the input level approaches 1.45 V. Out1 (also at 0 V), on the other hand, transitions to 4 V between 349 and 355 ns when the input level approaches 1.42 V. As before, the present invention maintains VIH/VIL trip point levels thereby guaranteeing a high level (4 V in this case) at Out2.

Sizing selected for PMOS and NMOS transistors should be such that sufficient drive is provided to OUT2 for a specific load, whether the output is a high or a low.

It is to be understood that although the present invention has been described with reference to a preferred embodiment, various modifications (such as NMOS and PMOS transistor ratios and sizing, varying circuit operating potentials or even developing a bipolar transistor version) known to those skilled in the art, may be made to the circuit presented herein without departing from the invention as recited in the several claims appended hereto.